1) Field of the Invention
The present invention relates to a method and a system for encoding and decoding telecommunication signals. In particular, the invention relates to a method and system that are useful in helping eliminate dynamic multi-path channel distortion problems common in fast fading wideband wireless communication environments, whilst simultaneously maintaining high data bandwidth transmission.
2) Description of Related Art
Many techniques for encoding and decoding telecommunications signals are known. One example is Orthogonal Frequency Division Multiplexing (OFDM), which is widely used in mobile communications, digital audio broadcasting and digital video broadcasting. OFDM provides a means of transmitting large data rates spread over multiple frequency channel sub-carriers, for example 52 (IEEE802.11a, g) or larger numbers such as 512, or 1024 carrier frequencies, see FIG. 1. Each frequency sub-carrier is modulated (e.g. PSK (Phase Shift Keying), QAM (Quadrature Amplitude Modulation) etc.) with 4, 5 or 6 bits of information, which over 1024 sub-carriers for example provides data rates of 4.096 Mbps, 5.12 Mbps and 6.144 Mbps respectively. Digital signal processing (DSP) techniques provide the means of producing the modulated multiple sub-carrier frequencies simply and efficiently rather than generating individual modulated sub-carrier frequencies and multiplexing them together.
FIG. 2 shows a very simplistic block diagram of an OFDM M-sub-carrier implementation. Here, M different bit pattern blocks of data, which consist of in-phase (I) and quadrature (Q) information, are presented to M inputs of an Inverse Fast Fourier Transform (IFFT) 10. This is the data to be modulated onto the M independent sub-carriers. The output components of the IFFT 10 are fed into a parallel-to-serial converter 12, which is clocked at the transfer rate of the system. This is followed by an anti-aliasing filter 14, after which the baseband modulated sub-carriers as shown in FIG. 1 are produced. It should be noted that sub-carrier 1 is modulated with the encoded bit patterns presented at input 1 of the IFFT; sub-carrier 2 with the encoded bit patterns presented at input 2 of the IFFT, etc.
Unfortunately, in the transmission of all these sub-carriers, sub-carrier channel distortion occurs due to signal echoes/fading or variations in attenuation between transmitter and receiver. This causes errors on decode as the original I and Q values are not reproduced at the receiver. Because these effects are frequency dependent, distortion influences differ across the frequency spectrum used by the sub-carriers. These channel distortion effects are often compensated for in OFDM through pilot symbol assisted schemes, which attempt to evaluate each channel's transfer function or distortion and compensate the received data accordingly. Basically, the pilot symbols are known modulated values, which are measured at the receiver and compared with the expected known true values. This allows the distortion effect of each sub-channel to be evaluated, removed and an estimate of the original signal of the sub-carrier recovered. These pilot symbols are embedded or interspersed with the data symbols, as shown in FIG. 3. For example, 4 pilot tone frequencies are interspersed across the 52 sub-carrier OFDM symbol transmission in the IEEE802.11a.
Estimates of the sub-carrier channel distortion of the data channels are made using the received values of the known pilot carrier channels. Estimates of the actual data carrying channel distortion effects are interpolated from the pilot channels, i.e. the actual channel effects of the data carrier sub-channels are not measured directly, but linearly interpolated from pilot channel determinations which are close to the data carrying sub-carrier channels themselves.
Two significant problems in implementing pilot carrier techniques exist. The first is achieving good signal-to-noise ratios for the pilot tones. Generally, the larger the strength of the pilot tones the better the channel distortion estimation. However, relatively large peak transmitted powers for the OFDM symbols can cause transmitter distortion. The problem is then choosing pilot symbols to minimise this distortion. Usually the variability of the peak power above the average value is measured in terms of the crest factor (CF). The crest factor can be minimised by designing pilot symbols, which have a random phase value. There are a number of techniques to assign such phase values, e.g. Shapiro-Rudin; Newmann; Narahasmi and Nojima algorithms. Adaptive optimisation techniques for minimum crest factors are also being considered.
The second problem with pilot carrier techniques relates particularly to communication systems in which channel response variations occur more rapidly across the wideband frequency spectrum of the subcarriers. In this situation, channel response effects between subcarrier frequencies can be poorly interrelated and a simple linear or non-linear extrapolation on current (or previous) pilot tone channel response evaluations is not sufficient. The time taken to recalibrate pilot tone channel coefficients to correct for distortion may result in valuable lost time, which could have been used for vital data transmission. In addition, for more radiply changing channel environments, once the new paradigm for the correction has been determined, the channel response may already have changed, resulting in the initiation of a further re-calibration of the pilot tones.
Recently a new method of modulation developed by T D Williams for use in mobile digital communication systems has been introduced, primarily to deal with the second problem presented above. This is described in U.S. Pat. No. 6,026,123. This is called Frequency Domain Reciprocal Modulation (FDRM). The main aim of this technique is to help eliminate dynamic multi-path channel distortion problems, which are common in wireless communications and thus provide a more robust and improved error rate communications link for OFDM systems. FDRM has many applications covering for example mobile telephony, mobile internet access, digital audio broadcasting, digital video broadcasting and microwave applications. FDRM is considered to be a companion to OFDM and so is implemented within OFDM type communication systems. FDRM can operate in a single carrier frequency modem, or in multi sub-carrier OFDM.
FDRM involves transmitting two packets or data blocks. These blocks are a normal block, which includes the data, and a reciprocal block, see FIG. 4. Each block could represent a symbol of M modulated sub-carriers in OFDM or indeed a single carrier frequency in a modem. In order to understand the basic principles of FDRM, a single sub-carrier frequency in OFDM will be considered. In FDRM, each of the normal and reciprocal transmission blocks contains the same data in a different way. For example, if the modulation technique for the sub-carrier is PSK then for a single sub-carrier the first data block is transmitted with amplitude A and phase angle φ, i.e. S1=A exp(jφ). This represents a digital block pattern transmission, which is determined by the amplitude and the phase angle on a constellation scatter diagram. Unfortunately, when an echo or fading signal is also received in a wireless communications link, the point in the received constellation diagram is rotated and the amplitude changes, see FIG. 5. This results in an error, because the decoded block pattern is now different from the original due its new position on the scatter plot.
Williams has shown, see U.S. Pat. No. 6,026,123 and IEEE Trans. on Broadcasting, Vol 45, pp. 11-15, March 1999 and Proc. 1999 NAB Broadcast Engineering Conference, Las Vegas, pp. 71-78, that for the time domain transmission, the effect of an echo signal can be expressed mathematically as follows:X(t)=S(t)+aS(t−T)where X(t) is the received signal, and S(t) the received signal when no noise or channel response is present. The term aS(t−T) is a received echo signal, and so causes channel distortion, with T equal to the delay time of the echo path. In the frequency domain, X(t) is transposed to:X(f)=S(f)(1+ae−j2πfT)This may be written as:X(f)=S(f)H(f)where H(f) is the complex frequency response associated with the transmission channel. The source of this distortion could include weak and strong echoes caused for example by moving mobile transmitters and/or receivers etc. If the transmission channel is perfect, then H(f) has an amplitude of one and a phase angle of zero, resulting in no rotation on the scatter plots.
To help alleviate the problems of channel distortion, FDRM transmits the original sub-carrier data block, followed immediately by the second data block, which contains the inverse of the original sub-carrier S i.e.:(1/A)exp(−jφ)
In the first instance, it is presumed that the distortion on the same channel has not altered significantly during the two block transmission. This would normally be correct as long as the data block transmission and propagation time is shorter than the dynamically varying effect of the channel. Letting the original transmitted sub-carrier be represented by S1 and the inverse sub-carrier be represented by S2, then after the same channel distortion (echo/fading) has affected both transmission sub-carriers, the received signals X1 and X2 respectively become:X1=S1H=A exp(jφ)H X2=S2H=(1/A)exp(−jφ)H A coherent quadrature detector measures the in-phase component (I) and the quadrature (90°) out-of-phase component (Q) of both transmitted signals. In this case, the received quadrature signals may be represented by:X1=I1+jQ1 X2=I2+jQ2 The solutions for recovering the originally sent, unimpaired, sub-carrier signal S, i.e. free from channel distortion and also attenuation, and also a measure of the channel response H are:
                    S        =                                            X              1                                      X              2                                                              H        =                                            X              1                        ⁢                          X              2                                          These can be re-expressed in terms of the I and Q components through:
      S    =                            (                                    I              1                        +                          jQ              1                                )                          (                                    I              2                        +                          jQ              2                                )                          H    =                            (                                    I              1                        +                          jQ              1                                )                ⁢                  (                                    I              2                        +                          jQ              2                                )                    It is easy to show mathematically, after a degree of manipulation of the equations, that with the inclusion of channel response, the recovered or estimated values of the undistorted original sub-carrier transmitted amplitudes can be determined as follows:
            A      ^        =                                        (                                          I                1                2                            +                              Q                1                2                                      )                                1            /            2                                                (                                          I                2                2                            +                              Q                2                2                                      )                                1            /            2                                          ϕ      ^        =                  1        2            ⁢                                    tan                          -              1                                ⁡                      [                                                                                I                    2                                    ⁢                                      Q                    1                                                  -                                                      I                    1                                    ⁢                                      Q                    2                                                                                                                    I                    1                                    ⁢                                      I                    2                                                  -                                                      I                    1                                    ⁢                                      Q                    2                                                                        ]                          .            An estimate of the recovered in-phase and quadrature components is given throughÎ=Â cos {circumflex over (φ)}{circumflex over (Q)}=Â sin {circumflex over (φ)}These are estimates of the original sub-carrier transmitted signals, not the received signals and thus the need for equalisers is not required. The gain of the transmitter (assuming it is constant), the attenuation over the transmission path, and also channel echoes have all been eliminated presuming of course the distortion has not altered significantly throughout the two block transmission. The amplitude of the sub-carrier is the normalised amplitude in relation to the definition of A=1 on the scatter plot. FDRM therefore has the capability to completely remove the effects of echo channel distortion and reproduce the original sub-carrier data free from multi-path echo signals. Of course, the inclusion of noise, or small variations due to a changing channel response, on the received detector quadrature components affects the error rate performance of these algorithms. These influences are discussed briefly later.
There are two major weaknesses with FDRM. Firstly, there are dramatic variations of the transmitted amplitude around the normalised amplitude value. When significant noise is present and the amplitude of S1 is increased to accommodate larger or smaller amplitude signals than the normalised value, for example those amplitude signals required in normal QAM modulation, then there are large errors on decoding. This arises due to the nature of the inverse amplitude of the sub-carriers being transmitted in the reciprocal data block transmission, i.e. the signal-to-noise of S2 decreases when the amplitude of S1, increases thus introducing decoding noise errors. For this reason a maximum normalised value of amplitude A=1.333 has been recommended for FDRM transmissions. In addition, it is believed that FDRM is only practically possible in modulation techniques that have no low energy frequency components or magnitudes. The best example of such a modulation technique is OFDM, where there is indeed an equal spread of energy frequency components. However, this cannot always be guaranteed in OFDM, particularly for QAM where the amplitudes can vary significantly.
Another disadvantage of FDRM is that due to the nature of the algorithms, and the inherent signs of the I and Q component values, the inverse tangent introduces a phase ambiguity of 180° for some of the decoded phase angle determinations. To eliminate this problem, pilot carrier tones within the transmissions have been suggested to track carrier phase changes with frequency in order to indicate the +ve in-phase (I) axis. However, this requires further information to be transmitted along with the data itself. Other solutions to this problem include the concepts of constellation scatter plots without 180° rotational symmetry, or small DC offsets in scatter points to locate the positive phase axis of the scatter diagram. These tend to increase the complexity of the technique, without providing significant overall improvements.